System, Method and Emulation Circuitry Useful For Adjusting a Characteristic of A Periodic Signal

ABSTRACT

Systems, methods and circuitry useful for adjusting a periodic signal such as with a voltage controlled oscillator or a delay line. In one series of embodiments, circuits and methods are provided for controlling current flow through first and second parallel paths where an impedance device in one path emulates the impedance characteristics of a different device in the other path. A phase or frequency characteristic of the periodic signal may be adjusted by alternate switching of current through the two paths.

PRIORITY BASED ON RELATED APPLICATION

This application claims priority based on U.S. Provisional ApplicationNo. 61/512,549 filed 28 Jul. 2011.

FIELD OF THE INVENTION

The present invention relates to electronic systems and, moreparticularly, to systems, methods and circuitry useful for adjusting aperiodic signal such as with a voltage controlled oscillator or a delayline. In one series of embodiments, circuits and methods are providedfor controlling current flow through first and second parallel pathswhere an impedance device in one path emulates the impedancecharacteristics of a different device in the other path. A phase orfrequency characteristic of the periodic signal may be adjusted byalternate switching of current through the two paths.

BACKGROUND

The phase locked loop (PLL) circuit is a feedback control circuit whichmay be analog or digital. A phase detector develops an adjustment signalbased on comparison between the output of a local voltage controlledoscillator (VCO) and a reference clock input signal. The adjustmentsignal is processed to provide a modified input to the VCO which resultsin a phase or frequency modification to the oscillator output signal.Phase locked loop circuits are common building blocks in customintegrated circuitry providing, for example, synchronization solutionsin a wide variety of gigaHertz (GHz) rate data communicationsapplications. However, in some applications, such as cellularcommunications base stations, high speed precision has required use ofdiscrete components.

Conventionally, PLL's can be categorized as analog or digital, butnumerous variants exist, including the combination of digital phasedetection with the phase detection output processed through a chargepump and an analog loop filter to provide a voltage input to the VCO. Asis well known, a charge pump comprises switches which control chargingof the capacitor in the loop filter to accumulate charge. See FIGS. 1. Afull digital PLL solution comprises a digital phase detector, a digitalfilter and a numerically controlled oscillator. Both analog and digitalimplementations typically generate a proportional component and anintegral component for, respectively, delivering phase and frequencyfeedback control to the oscillator.

FIG. 1A is a high level diagram of a conventional PLL incorporating acharge pump an analog loop filter and a trans-conductance (G_(m))amplifier as more fully illustrated in FIG. 1B. A phase-frequencydetector (PFD) receives a reference clock input signal of desiredfrequency and a feedback signal from a VCO. The PFD may be one ofnumerous designs, including types based on exclusive OR gates or flipflops, which output a pulse signal proportional to positive or negativephase and frequency differences between the clock signal and thefeedback signal.

In the past, it has been necessary to provide the charge pump, loopfilter, and G_(m) amplifier to translate the full swing up and downsignals from the PFD to the VCO. The signals from the PFD turn switchesin the charge pump on and off to provide currents, creating a voltagedifferential, ΔV, across the resistor R. This small signal voltage, ΔV,is then passed into the G_(m) amplifier, sometimes referred to as avoltage to current converter. The current output from the G_(m) goesinto the VCO. Through this process, gain is realized through the productof the charge pump current and resistance of R, as well as operation ofthe G_(m) amplifier. However, the analog PLL requires a large passivedevice in a monolithic process and results in a path for noise to enterthe VCO.

The VCO may be a three stage ring oscillator circuit having threeinverters I₁, I₂, I₃ coupled in series as shown in FIG. 1C. Assuming apredetermined bias voltage, the circuit oscillates at a frequency, f,having an associated period 1/f. For this three stage ring, the groupdelay (or the phase shift) of all 3 stages is 360 degrees. Hence, thismeans each stage I₁, I₂, I₃ has a delay of 120 degrees and, due to thephase shift, the nodes labeled N₁, N₂, N₃ will be at differentpotentials at any instant in time. For example, when one of the nodes isclose to V_(DD), another node will be close to ground (V_(SS)) and theother node will be at a potential between V_(DD) and ground.

Generally, the desired VCO frequency, f, is a multiple, N, of thereference clock signal frequency, and is factored by the block DIV/Naccordingly to provide the suitable feedback signal for comparison to bemade by the PFD. This results in a phase difference output signal whichmay comprise a pulse width having a time duration in proportion to thephase difference. A charge pump receives the phase difference outputsignal and generates current in proportion to the phase difference. Thecurrent output by the charge pump is fed through an analog loop filterto the VCO. The design of the loop filter affects response time,bandwidth and stability. The combination of the charge pump and the loopfilter provides two components of signal to the VCO: a pulse componentin proportion to the phase difference, and an integral component whichaffects the frequency adjustment.

An advantage of the analog PLL is low jitter. However, with increaseddemands for higher speed precision, even the relatively low noise analogPLL implementations considered acceptable at lower data rates may be toonoise sensitive in some gigahertz data communications. For example,tuning of the loop filter component for a desired response time andstability in an analog PLL can still result in added noise. Generally,it is desirable to develop designs which further reduce the impact ofnoise sources. Another limitation of analog PLL circuits is that theanalog charge pumps and loop filters have wide range voltage tuningrequirements. These are becoming increasingly difficult to meet asmanufacturing technologies have moved past the 45 nanometer node to 28nm technologies and toward, for example, 10 nm linewidths. Whenfabricating analog PLLs in deep nanometer technologies there are alsoconcerns about relatively high capacitor leakage rates and, generally,disadvantages due to an inability to scale the sizes of analogcomponents with the smaller digital components.

FIG. 2 illustrates an example of an all digital PLL. Common to all fullydigital PLLs, analog circuit blocks are replaced by converting signalsreceived from the PFD into digital signals, using quantizers oranalog-to-digital converters. In lieu of a charge pump and an analogloop filter, the digital implementation performs digital conversions ofthe output signals generated by the PFD. Elimination of the capacitorpermits better scaling to small fabrication geometries and reducessensitivity to process variations. The illustrated digital PLL has aproportional path, for adjusting the phase of the VCO, which is distinctfrom a frequency adjusting integral path. The proportional and integralpaths undergo separate digital-to-analog conversions for input to theVCO because they each may require conversion of a different number ofbits. Advantageously, elimination of the analog charge pump and analogloop filter enhances scalability and avoids sensitivity problems whichanalog components exhibit to minor process variations. On the otherhand, quantization of the proportional and integral tuning pathsintroduces jitter, e.g., static phase offset, which precludes use ofdigital PLLs when timing precision is essential.

SUMMARY OF THE INVENTION

In one series of embodiments according to the invention, a method isprovided for changing the phase or frequency of a first periodic signalwith respect to a second periodic signal, the first periodic signalbeing output from a first device having a nonlinear impedancecharacteristic in a voltage operating range. Discrete control signaltypes of variable time width are provided where the time width is inproportion to a phase difference between the first and second signals. Afirst of the signal types is indicative of a negative difference inphase or frequency of the first periodic signal relative to the phase orfrequency of the second periodic signal. A second of the signal types isindicative of a positive difference in phase or frequency of the firstperiodic signal relative to the phase or frequency of the secondperiodic signal. Control signals of the first type are periodicallyapplied to a first switching device to control current flow along afirst circuit path, from a voltage source through the first switchingdevice, through the first nonlinear device and to a reference voltageterminal, to adjust a phase or frequency characteristic of the firstsignal output from the first nonlinear device. Signals of the secondtype are periodically applied to second switching circuitry to controlcurrent flow along a second circuit path in parallel with the firstcircuit path from the voltage source through the second switchingdevice, through a second impedance device having a nonlinear impedancecharacteristic and to the reference voltage terminal. The first andsecond switching devices are operated in to response to variations inthe signals of the first and second types to switch current flow betweenthe first and second circuit paths so that at times current flows onlythrough the first circuit path and then only through the second circuitpath. Impedance characteristics of the first nonlinear device and thesecond nonlinear impedance device are so matched that current-voltagecharacteristics of the first and second impedance devices are limited,e.g., within ten percent of one another, advantageously within fivepercent of one another or in a range less than said five percent (e.g.,less than two percent of one another) throughout the voltage operatingrange of the VCO, and the current-voltage characteristics of the firstand second impedance devices may be perfectly matched at least onevoltage or current level or across an entire operating range.

Also according to an embodiment of the invention a circuit is providedwhich is suitable for tracking impedance characteristics of anoscillator. The circuit includes a first PMOS FET, a second PMOS FET, afirst NMOS FET and a second NMOS FET. The first PMOS FET has a gateregion connected to receive a reference voltage. The second NMOS FET 108has a gate region connected to receive a supply voltage level. Thesecond PMOS FET 104 has a gate region connected to a node between thefirst and second NMOS FETs 106 and 108. The first NMOS FET has a gateregion connected to a node between the first and second PMOS FETs 102and 104. In one implementation the oscillator is a three stage inverterring oscillator and the first PMOS FET and the second NMOS FET operatein a triode mode, which corresponds to the impedances of FET transistorsin the oscillator that are fully switched when the associated gate biasvoltage is at the supply voltage level or the reference level while thesecond PMOS FET and the first NMOS FET operate in the saturation modewhich corresponds to the impedances of FET transistors in the oscillatorthat are partially switched on when associated gates are biased at avoltage between the supply voltage level and the reference level.

BRIEF DESCRIPTION OF THE DRAWINGS

Features of the invention will be best understood when the followingdetailed description is read in conjunction with the accompanyingdrawings wherein:

FIG. 1A illustrates an analog phase locked loop circuit incorporating acharge pump, an analog loop filter and a trans-conductance (G_(m))amplifier;

FIG. 1B further illustrates components of the circuit shown in FIG. 1A;

FIG. 1C illustrates a conventional ring oscillator circuit;

FIG. 2 FIG. 2 illustrates a digital phase locked loop circuit;

FIG. 3 illustrates a phase locked loop circuit according to anembodiment of the invention;

FIG. 4 illustrates exemplary quantizer circuitry for the embodimentshown in FIG. 3;

FIG. 5 illustrates exemplary processing circuitry, including anaccumulator and a sigma delta modulator, for the embodiment shown inFIG. 3;

FIG. 6A illustrates exemplary control and interface circuitry for theembodiment shown in FIG. 3, including three paths of control, eachproviding an adjustment signal;

FIG. 6B illustrates placement of power supply rejection circuitry inrelation to the control and interface circuitry of FIG. 6A;

FIGS. 7A-7C illustrate exemplary timing diagrams for an illustrativesingle ended implementation of proportional path circuitry shown in FIG.6 to control the timing of operation of the switches;

FIG. 8 illustrates an exemplary differential implementation ofproportional path circuitry which may be applied in lieu of the singleended implementation of proportional path circuitry shown in FIG. 6;

FIGS. 9A-9C are timing diagrams illustrating operation of controlsignals and current flow in the differential implementation of theproportional path circuitry shown in FIG. 8;

FIG. 10 illustrates an embodiment of a replica circuit 65 havingimpedance characteristics which suitably match the characteristics of aVCO;

FIG. 11 provides a graphic comparison between voltage-currentcharacteristics of the replica circuit of FIG. 10 and voltage-currentcharacteristics of a VCO;

FIGS. 12-14 illustrate example embodiments of the power supply rejectioncircuitry of FIG. 6B; and

FIG. 15 is a simplified schematic illustration ofserializer/deserializer components incorporating phase locked loopcircuits according to the invention.

Like reference numbers are used throughout the figures to denote likecomponents. Numerous components are illustrated schematically, it beingunderstood that various details, connections and components of anapparent nature are not shown in order to emphasize features of theinvention. Various features shown in the figures are not shown to scalein order to emphasize features of the invention.

DETAILED DESCRIPTION

FIG. 3 illustrates a phase locked loop (PLL) circuit 10 according to anembodiment of the invention. A voltage controlled oscillator (VCO) 12outputs a signal 14 at a terminal 15 thereof, the phase and frequency ofwhich are adjustable based on a comparison between the signal 14 and aclock reference signal 16. The VCO 12 may be the ring oscillator circuitshown in FIG. 1C. In this example, the desired frequency of the signal14 is a multiple N of the reference clock signal frequency. To effectphase and frequency control of the VCO 12, a portion of the outputsignal 14 is factored with divide by N circuitry (DIV/N) 17 to provide afeedback signal 18 as an input to a terminal 19 of a Phase-FrequencyDetector (PFD) 20. The reference clock signal 16 is provided as an inputto a terminal 21 of the PFD 20.

Control and interface circuitry 22, positioned between the VCO 12 andthe PFD 20, receives inputs both directly and indirectly from the outputsignal 24 of the PFD 20, which is in analog form, to provide the VCO 12with a combination of input signals. In the illustrated embodiment theinput signals to the control and interface circuitry 22 are acombination of the analog version (i.e., the output signal 24) and adigitized version of the output signal 24. Based on the combination ofthe control and interface.

Operation of the circuitry 22 is controlled with both an analog versionand a digital version of the PFD output signal 24 to provide multipleinput adjustment signals 30 to the VCO 12.

As illustrated in FIG. 6A, the circuitry 22 receives the output signal24 in analog form directly from the PFD 20. This analog signal controlsswitches in proportional path circuitry to provide first adjustmentsignals (e.g., current signals) to the VCO 12 that contribute toadjustment of the phase of the VCO output signal. In the illustratedembodiment, the circuitry 22 also receives a digitized version of thefirst analog VCO input signals indirectly from the PFD 20. This digitalsignal generates a level of current injection in the circuitry 22 thatcontrols other current signals sent to the VCO, i.e., through integralpath circuitry in addition to the proportional path circuitry. In theillustrated embodiment the digital signal generates a signal (isum)which controls current sent to the VCO 12 through fast integral pathcircuitry and through slow integral path circuitry. In response to thecontrol signal (isum) the fast integral path circuitry provides a secondadjustment signal through a fast integral path to the first inputterminal of the VCO. In response to a filtered version of the controlsignal (fisum), the slow integral path circuitry provides a thirdadjustment signal through a fast integral path to the first inputterminal of the VCO.

The digital signal also provides a level of control to the currentsignals sent to the VCO 12 through the proportional path circuitry.Collectively, these individual signals, referenced in FIG. 3 as acombined signal 30, adjust the phase and frequency of the VCO 12relative to the reference signal 16.

The PFD 20 generates analog signals UP, DN, UN, DP referred to herein asthe signals 24, which are indicative of a phase-frequency differencebetween the VCO feedback signal 18 and the clock reference signal 16.The signal UP indicates that an increase in voltage input to the VCOwill reduce a phase-frequency difference between the reference clocksignal frequency and the feedback signal 18. The signal DN indicatesthat a decrease in voltage input to the VCO will reduce aphase-frequency difference between the reference clock signal frequencyand the feedback signal 18. The signal UN is the inverse of the signalUP and the signal DP is the inverse of the signal DN. The signals 24 areprovided as a first input, fed directly from the PFD 20, to the controland interface circuitry 22.

Given that the PFD provides a signal 24 every reference clock period,its output signals, UP and DP, are discretized. These signals 24 fromthe PFD 20 are also fed to quantizer circuitry 34, such as illustratedin FIG. 4 comprising logic circuitry which quantizes the signals 24,thereby providing as an output a series of digital adjustment signals 36comprising UPINTN, UPINTP, DNINTP and DNINTN. UPINTP is the complementof UPINTN and DNINTN is the complement of DNINTP.

The adjustment signals 36 are further processed via digital signalprocessing circuitry 40 to provide an M bit wide second input signal 38to the control and interface circuitry 22. With reference also to FIG.5, the processing circuitry 40 comprises an accumulator 42 and a sigmadelta modulator 44. The accumulator 42 is programmable based onalgorithm inputs as shown in FIG. 5. In the illustrated embodiment, thequantizer circuitry 34 and the accumulator 42 operate under the controlof a clock signal CLKACC which, in this example, is the same as thereference signal 16, but the clock signal input to the accumulator 42may be at a different frequency than the reference clock signal. Theaccumulator receives a series of high resolution, e.g., 22-bit wide,adjustment signals 36, UPINTP and DNINTP, at, for example, a rate of 100MHz. The modulator 24 operates under a clock signal CLKMOD which may,for example, be three times the frequency of the signal CLKACC.

The accumulator 42 accumulates 22 bit values based on a differencebetween the adjustment signals 36, e.g., (UPINTP−DNINTP), received fromthe quantizer 34, with a programmable gain, and up to 22-bit resolution.The accumulator 42 performs a function equivalent to the analogcharge-pump and capacitor:

ACC[n]=ACC[n−1]+GAIN*(UPINTP−DNINTP)

for n samples. The variable GAIN controls the speed at which theaccumulator accumulates. A high GAIN value allows the accumulator 42 toaccumulate faster. However, a high GAIN value also introduces morejitter due to increased integral loop gain which degrades stability. Alow GAIN value allows the accumulator to accumulate at a slower rate. Alower GAIN value also decreases the integral loop gain, rendering thePLL more stable. Taking advantage of these conditions, different modesof operation are defined and the optimum GAIN value can be chosen for atleast three modes: start up condition, normal operating mode, andspecial conditions for dynamic modes of operation.

During initial start up of the PLL circuit 10, the frequency of the VCO12 is adjusted in open loop mode to a frequency very close to thedesired frequency. The term open loop mode as used herein refers to astate in which the PLL circuit 10 is not in a closed loop mode.Typically, the difference between the adjusted value in the open loopmode and the desired frequency is 0.5% to 1% of the desired frequency.Once the adjustment is made, the loop is closed so that the PLL goesinto its locking mode. The integral loop then reacts to compensate forthe 0.5% to 1% frequency offset to ensure the PLL circuit achieves afinal VCO frequency equal to the desired frequency. Since theaccumulator is a 22-bit word, accumulating one bit at a time per clockcycle is a very slow process as confirmed by the above equation forACC[n]. In order to accelerate the locking process, an algorithm isprovided that modulates the GAIN in a logarithmic fashion for each timeperiod. In one embodiment the GAIN starts at a very high value, forexample, 2¹². For each programmable time delay, e.g., one microseceond,that lapses, the GAIN becomes the previous GAIN value divided by 2. Thistime delay is programmed from a preset 4-bit register, having a range of1 to 16 microseconds.

GAIN=GAIN/2.

This reduction in GAIN stops when the GAIN equals 8. Using this method,the PLL circuit 10 achieves frequency locking very quickly while theproportional path ensures that the VCO output signal 14 also stays phaselocked.

During normal operating mode, the register GAIN is maintained at a fixedvalue, e.g., 2³. Special Conditions for Dynamic Modes of Operation occurwhenever an external condition disturbs the loop, causing a frequencyshift in the VCO output signal 14. The PLL must react quickly to recoverfrom such disturbances and return to frequency and phase locked.However, as mentioned above, the integral loop has a relatively slowresponse time and may require a relatively long period of time in orderfor the PLL circuit 10 to recover from disturbances.

According to an embodiment of the invention, upon detecting repetitivecycles of consecutive UPINTPs, or upon detecting repetitive cycles ofconsecutive DNINTPs, the value of the GAIN is changed based onpre-established criteria. The criteria may establish a threshold numberof cycles during which there are only consecutive signals UPINTP or onlyconsecutive signals DNINTP upon which occurrence the GAIN is changed inaccord with a program. Hence, the GAIN changes dynamically. For example,with a normal mode of operation occurring over a given time period, theGAIN might be 2³. If a threshold algorithm determines that a counter,which only counts consecutive occurrences of the same signal, hasreached a threshold number (e.g., eight, corresponding to an occurrenceof eight consecutive signals UPINTP, or eight consecutive signalsDNINTP), then a dynamic change in the GAIN is triggered.

More specifically, if, after an external condition disturbs the PLLcircuit 10, a string of consecutive signals UPINTP is received by theaccumulator, i.e., with no signal DNINTP between any signal UPINTP, aGAIN change event is triggered in accord with the following conditionalstatement:

If (consecutive UPINTP or consecutive DNINTP is met) then GAIN=GAIN*2

As soon as the GAIN value is changed, the threshold algorithm resets thecounter to zero and the count re-starts upon occurrence of twoconsecutive values of the same signal (e.g., UPINTP or DNINTP). However,if there are no consecutive ups or downs for a predeterminedprogrammable time, then the GAIN is reset in accord with

GAIN=GAIN/2

thereby reverting to the GAIN value applied during the normal mode ofoperation, e.g., 2³. Also, each time the counter is incremented abovethe zero value but stops because two or more consecutive values of thesame signal (e.g., UPINTP) are followed by a different signal (e.g.,DNINTP), the counter is reset.

The modulator 44 applies a pulse density technique to translate thedigital adjustment signals 36 into a series of lower resolution words(e.g., M=8 bits) at a higher clock rate (e.g., with the frequency ofCLKMOD set to 600 MHz), as a second input signal 38 to the control andinterface circuitry 22. The signal 38 is applied to adjust the VCO inputsignal 30. Summarily, the second input signal 38 modifies the frequencyof the VCO output signal 14 relative to the reference signal 16 whilethe first input signal, i.e., the portion of the signal 24 fed directlyto the control and interface circuitry 22, modifies the phase of the VCOoutput signal 14.

A feature of embodiments of the invention is that the signals 30provided by the control and interface circuitry 22 comprise threecomponents, each generated via one of three different paths of control:a proportional circuit path, a fast integral circuit path and a slowintegral circuit path. This functionality is schematically illustratedin FIG. 6A. The circuitry 22 includes current source signal drivingcircuitry, also referred to herein as control circuitry 47, whichprovides a control signal isum to each of three paths leading to aninput node 45 of the VCO 12. The signal isum is delivered to circuitry46 which forms the fast integral circuit path. The signal isum thenpasses through a low pass filter 49 to provide a signal V_(bias). Thesignal V_(bias) controls current passing through circuitry 48 whichforms the slow integral circuit path. The signal V_(bias) is also fed tothe circuitry 60, 60′ of the proportional circuit path. See, also, FIG.8.

The control circuitry 47 is connected between a supply voltage railV_(DD) and a reference or ground rail V_(SS). A diode is formed with aFET 50 having its drain 52 tied to the gate 54. A low pass filtercapacitor 58 is connected between the gate 54 and the source 56 oftransistor 50. Two digital-to-analog converters (DACs) 62, 64 areconnected in parallel between the drain 52 and V_(SS).

In the example embodiment the DAC 62 provides a steady state currentinjection which is programmable based on a four bit input (M=4). Thecurrent level from the DAC 62 is set during an initial calibration ofthe VCO output signal 14. The DAC 64 receives the eight bit (M=8) signal38 from the Sigma Delta modulator 44 which, for example, may be fed inat a 600 MHz clock frequency to modulate the gate voltage signal isum,which directly or indirectly controls signals fed to the input node 45from each of the three circuit paths.

Proportional Circuit Path

Functional implementation of the proportional circuit path isillustrated with a single ended implementation as shown with thecircuitry 60 in FIG. 6A. FIG. 8 illustrates an exemplary differentialimplementation of the proportional path circuitry, indicated ascircuitry 60′. The circuitry 60, 60′ eliminates the need for including acharge pump and an analog loop filter in the proportional path.

The proportional path circuitry 60 receives the pulse signals UP and DPhaving variable time widths from the PFD 20. The difference in pulsewidth between UP and DP is proportional to the phase difference betweenthe clock reference signal 16 and the feedback signal 18. Theproportional path circuitry 60, connected between V_(DD) and V_(SS), iscontrolled to provide only a proportional path through the input node 45to the VCO 12, or only a path through a replica circuit 65 havingimpedance characteristics which closely follow those of the VCO 12, orno current flow through either the VCO 12 or the replica circuit 65. Thereplica circuit 65 is positioned between the node V_(REP) and V_(SS).Current flow through one or the other paths, i.e., to the VCO 12 or tothe replica circuit 65, is determined by switches 64 or 66 in eachbranch. Operation of the switch 64 is controlled by the signal UP. Whenthe signal UP goes from a low level to a high level the switch 64 isclosed. Otherwise the switch 64 is open. Similarly, operation of theswitch 66 is controlled by the signal DN. When the signal DN goes from alow level to a high level the switch 66 is closed. Otherwise the switch66 is open.

A current mirror is set up in the proportional path with a FET 68. Withreference to the timing diagrams of FIGS. 7A-7C, the relative delaybetween the signals UP and DP and the difference in pulse width betweenthe signals UP and DP control the timing of operation of the switches 64and 66.

The signal UP transitions from a logic low level to a logic high levelin accord with transitions of the reference clock signal 16 from a lowvoltage level to a high voltage level, e.g., at fifty percent of thehigh voltage level. Similarly, the signal DP transitions from a logiclow level to a logic high level in accord with transitions of thefeedback clock signal 18 from a low voltage level to a high voltagelevel, e.g., at fifty percent of the high voltage level.

The switch 64 transitions from an open position to a closed positionwhen the signal UP transitions from a logic low voltage level to a logichigh voltage level; and the switch 66 transitions from an open positionto a closed position when the signal DP transitions from a logic lowvoltage level to a logic high voltage level.

With reference to FIG. 7A, when the phase of the reference clock signal16 leads the phase of the feedback clock signal 18 by a time Δt₁, thesignal UP transitions from a logic low voltage level to a logic highvoltage level a time Δt₁ before the signal DP transitions from a logiclow voltage level to a logic high voltage level. This causes the switch64 to transition from an open configuration to a closed configuration atime Δt₁ before the signal DP transitions from a logic low voltage levelto a logic high voltage level. When the feedback clock signal 18transitions from a low voltage level to a high voltage level, causingthe signal DP to transition from a logic low voltage level to a logichigh voltage level, both the signal UP and the signal DP then transitionfrom a logic high voltage level to a logic low voltage level therebyplacing each of the switches 64, 66 in an open position, terminatingcurrent flow through the proportional path circuitry 60.

With reference to FIG. 7B, when the phase of the reference clock signal16 lags the phase of the feedback clock signal 18 by a time Δt₂, thesignal DP transitions from a logic low voltage level to a logic highvoltage level a time Δt₂ before the signal UP transitions from a logiclow voltage level to a logic high voltage level. This causes the switch66 to transition from an open configuration to a closed configuration atime Δt₂ before the signal UP transitions from a logic low voltage levelto a logic high voltage level. When the reference clock signal 16transitions from a low voltage level to a high voltage level, causingthe signal up to transition from a logic low voltage level to a logichigh voltage level, both the signal UP and the signal DP then transitionfrom a logic high voltage level to a logic low voltage level therebyplacing each of the switches 64, 66 in an open position, terminatingcurrent flow through the proportional path circuitry 60.

With reference to FIG. 7C, when there is no phase difference between thereference clock signal and the feedback clock signal 18, the signals UPand DP simultaneously transition from a logic low voltage level to alogic high voltage level, whereby both of the switches 64, 66 are heldin an open position, preventing any current flow through theproportional path circuitry 60. The signals UP and DP thensimultaneously transition from a logic high voltage level to a logic lowvoltage level with the switches 64, 66 still remaining in openconfigurations such that no current flows through the proportional pathcircuitry 60.

Next, referring to FIG. 8, the circuitry 60′ includes first and secondPMOS FETs 80, 82, each acting as a current mirror in one of two crosscoupled branches 84, 86 connected between V_(DD) and V_(SS), in a manneranalogous to the circuitry 60. In this example, the source of FET 80 isconnected to V_(DD) and the drain of FET 80 is connected to the sourceof each of two PMOS FETS 90, 92. The drain of FET 90 is connected as aninput to the replica circuit 65 and the drain of FET 92 is connected asan input to the VCO 12. The source of FET 82 is connected to V_(DD) andthe drain of FET 82 is connected to the source of each of two PMOS FETS96, 98. The drain of FET 96 is connected as an input to the VCO 12 andthe drain of FET 92 is connected as an input to the replica circuit 65.

The gate of FET 90 receives the control signal UP as described withrespect to the circuitry 60, and the gate of FET 98 receives the controlsignal DN, also as described with respect to the circuitry 60. The gateof FET 92 receives a control signal UN, which is the complement of thecontrol signal UP, and the gate of FET 96 receives the control signalDP, which is the complement of the control signal DN.

FIGS. 9 are timing diagrams illustrating operation of the controlsignals in the differential implementation of the proportional circuitpath based on the circuitry 60′ and the resulting current flow throughthe proportional circuit path into the VCO 12. Since the control signalsUN and DN are complements of UP and DP, respectively, only UP and DP areexpressly shown in FIGS. 9.

FIG. 9A illustrates logic levels of the control signals and resultingcurrent flow to the VCO 12 when, as described with reference to FIG. 7A,the phase of the reference clock signal 16 leads the phase of thefeedback clock signal 18. With the phase of the reference clock signal16 leading the phase of the feedback clock signal 18 by the time Δt₁,the phase detector 20 transitions the control signal UP from a logic lowvoltage level to a logic high voltage level. This places FET 90 in anon-conducting mode, preventing current flow from FET 80 into thereplica circuit 65. At the same time that the signal UP transitions tothe logic high voltage level, the signal UN transitions from a logichigh voltage level to a logic low voltage level. This places FET 92 in aconducting mode, sending current from FET 80 into the VCO12 during thetime interval Δt₁. Prior to and during the time interval Δt₁ (i.e.,before the feedback clock signal 18, shown in FIG. 7A, transitions froma low voltage level to a high voltage level) the control signal DP is ata logic low voltage level and the complement DN is at a logic highvoltage level. Thus, prior to and during the time that FET 92 is in aconducting state, FET 96 is also in a conducting state, sending currentfrom FET 82 into the VCO12. Also, while FETs 92 and 96 are in conductingstates, FET 98 is in a non-conducting mode, preventing current flow fromFET 82 into the replica circuit 65. Consequently, during the timeinterval Δt₁, both of the branches 84 and 86 feed current to the VCO 12while the replica circuit 65 receives no current. Thus the current intothe VCO increases from a steady state level to a higher level during theperiod Δt₁.

Once the time period Δt₁ lapses, the feedback clock signal 18transitions from a low voltage level to a high voltage level, such thatthe phase detector 20 transitions the control signal DP from a logic lowvoltage level to a logic high voltage level, after which time the phasedetector transitions both the signal UP and the signal DP from logichigh voltage levels to logic low voltage levels. Simultaneously,complements of each, UN and DN, transition from logic low voltage levelsto logic high voltage levels. With the signals UP and DP at logic lowvoltage levels and the signals UN and DN at logic high voltage levels,the VCO receives the steady state current level through FET 96 only andthe replica circuit receives a similar current level through FET 90only.

FIG. 9B illustrates logic levels of the control signals and resultingcurrent flow to the VCO 12 when, as described with reference to FIG. 7B,the phase of the reference clock signal 16 lags the phase of thefeedback clock signal 18. With the phase of the reference clock signal16 lagging the phase of the feedback clock signal 18 by the time Δt₂,the phase detector 20 transitions the control signal DP from a logic lowvoltage level to a logic high voltage level. This places FET 96 in anon-conducting mode, preventing current flow from FET 82 into the VCO12. At the same time that the signal DP transitions to the logic highvoltage level, the signal DN transitions from a logic high voltage levelto a logic low voltage level. This places FET 98 in a conducting mode,sending current from FET 82 into the replica circuit 12 during the timeinterval Δt₂. Prior to and during the time interval Δt₂ (i.e., beforethe reference clock signal 16, shown in FIG. 7B, transitions from a lowvoltage level to a high voltage level) the control signal UP is at alogic low voltage level and the complement UN is at a logic high voltagelevel. Thus, prior to and during the time that FET 98 is in a conductingstate, FET 90 is also in a conducting state, sending current from FET 80into the replica circuit 65. Also, while FETs 90 and 98 are inconducting states, FET 92 is in a non-conducting mode, preventingcurrent flow from FET 80, through FET 92 and into the VCO 12.

Consequently, with the phase of the reference clock signal 16 laggingthe phase of the feedback clock signal 18, during the time interval Δt₂,both of the branches 84 and 86 feed current to the replica circuit 65while the VCO 12 receives no current from either of the branches 84, 86.Thus the current into the VCO decreases from a steady state level to alower level during the period Δt₂. Once the time period Δt₂ lapses, thereference clock signal 16 transitions from a low voltage level to a highvoltage level, such that the phase detector 20 transitions the controlsignal UP from a logic low voltage level to a logic high voltage level,after which time the phase detector transitions both the signal UP andthe signal DP from logic high voltage levels to logic low voltagelevels. Simultaneously, complements of each, UN and DN, transition fromlogic low voltage levels to logic high voltage levels. With the signalsUP and DP at logic low voltage levels and the signals UN and DN at logichigh voltage levels, the VCO again receives the steady state currentlevel through FET 96 only and the replica circuit receives a similarcurrent level through FET 90 only.

FIG. 9C illustrates logic levels of the control signals and resultingcurrent flow to the VCO 12 when, as described with reference to FIG. 7C,there is no phase difference between the reference clock signal 16 andthe feedback clock signal 18. When simultaneously receiving the leadingedges of the reference clock signal 16 and the feedback clock signal 18,the phase detector 20 transitions both of the control signals UP and DPfrom a logic low voltage level to a logic high voltage level. Thisplaces FETs 90 and 96 in a non-conducting mode, preventing current flowfrom FET 80 into the replica circuit 65 and preventing current flow fromFET 82 into the VCO 12. Simultaneous with the transitions of both thecontrol signals UP and DP to logic high voltage levels, the controlsignals UN and DN both transition from a logic high voltage level to alogic low voltage level. This places FETs 92 and 98 into conduction,resulting in passage of current from FET 80 into the VCO 12 and passageof current from FET 82 into the replica circuit 65. Thus there is anexchange in current flows from the steady state arrangement, where FET92 feeds current to the VCO and FET 98 feeds current to the replicacircuit, to an arrangement which lasts a relatively short period whereFET 96 feeds current to the VCO and FET 90 feeds current to the replicacircuit. The phase detector then transitions the control signals back tothe values corresponding to the steady state condition where UP and DPare at logic low voltage levels and UN and DN are at logic high voltagelevels such that the VCO again receives the steady state current levelthrough FET 96 only and the replica circuit receives a similar currentlevel through FET 90 only.

With further reference to FIGS. 8 and 9, in the absence of the pulses UPand DP, half of the current flows into the replica circuit 65 and halfof the current flows through the VCO 12. Two paths of current flow fromthe first and second PMOS FETs 80, 82. In the absence of the pulses UPand DP, current flows along a first path from the FET 80 through the FET90 to the replica circuit 65 and current flows along a second path fromthe FET 82 to the FET 96 to the VCO 12. With reference to FIG. 9C, oncethe pulses UP and DP are issued, current flows along one path from theFET 80 through the FET 92 to the VCO 12 and along a second path from theFET 82 through the FET 98 to the replica circuit 65.

With reference to FIG. 9A, when the signal UP leads the signal DP, whenUP initially rises to a voltage level high (i.e., while the signal DP isstill at a voltage level low) current flows along a first path from theFET 80 through the FET 92 to the VCO 12 and along a second path from theFET 82 through the FET 96 to the VCO 12 such that the VCO receives twicethe current relative to the current received prior to the signal UPhaving reached a logic high. When the signal DP also rises to a voltagelevel high, the current paths are the same as described with referenceto FIG. 9C, i.e., once the pulses UP and DP are issued, current flowsalong one path from the FET 80 through the FET 92 to the VCO 12 andalong a second path from the FET 82 through the FET 98 to the replicacircuit 65.

With reference to FIG. 9C, when the signal DP leads the signal UP, whenDP initially rises to a voltage level high (i.e., while the signal DP isstill at a voltage level low) current flows along a first path from theFET 82 through the FET 98 to the replica circuit 65 and along a secondpath from the FET 80 through the FET 90 to the replica circuit 65 suchthat the VCO 12 receives no current from the proportional path while thereplica circuit 65 receives twice the current relative to the currentreceived prior to the signal DP having reached a logic high. When thesignal UP also rises to a voltage level high, the current paths are thesame as described with reference to FIG. 9C, i.e., once the pulses UPand DP are issued, current flows along one path from the FET 80 throughthe FET 92 to the VCO 12 and along a second path from the FET 82 throughthe FET 98 to the replica circuit 65.

Operation of the proportional path circuitry 60′ according to theillustrated examples of FIGS. 9 is characterized by relatively smoothtransitions as different FETs simultaneously switch in and out ofconduction. In the example illustration of circuitry 60′, all of thetransistors 90, 92, 96 and 98 have identical characteristics. Moregenerally, in other embodiments, the FETs 92, 96 and the FETs 90, 98 arematched pairs.

In the illustrated proportional path circuitry, stabilization of thevoltage input to the replica circuit 65, with respect to the voltageinput to the VCO 12, is not based on feedback. Instead, to minimizeinjection of transient glitches into the VCO during operation of theproportional path circuitry, the replica circuit 65 closely matches thevoltage-impedance characteristics of the VCO 12. An embodiment of areplica circuit 65 having impedance characteristics which suitably matchthe characteristics of the VCO 12 is shown in FIG. 10. Recognizing thatthe VCO 12 is a non-linear device, the replica circuit 65 is a DCcircuit which emulates the input impedance characteristics of the VCO.That is, when the input into the VCO 12 changes, the impedance of theVCO also changes. For a predefined or characteristic operating range ofthe VCO 12 in the phase locked loop circuit 10, FIG. 11 illustrates howthe voltage-current characteristics V_(REP) of the replica circuit 65track the voltage-current characteristics V_(OSC) of the VCO 12. Withthe phase locked loop (PLL) circuit 10 not incorporating charge pumpcircuitry to store charge in a capacitor for input to the voltagecontrolled oscillator, the impedance characteristics of the replicacircuit 65 and the VCO 12 are so matched that current-voltagecharacteristics of these two impedance devices can be within fivepercent of one another throughout the voltage operating range of theVCO. With substantially matched impedance characteristics of the VCO 12and the replica circuit 65, the system does not incorporate charge pumpcircuitry to store charge in a capacitor for input to the voltagecontrolled oscillator.

The impedance characteristics of the replica circuit 65 can so closelyfollow the impedance characteristics of the voltage controlledoscillator as a function of voltage level as to allow a voltage level tobe switched between the subcircuit 65 and the voltage controlledoscillator 12 without creating voltage spikes when a voltage level isswitched between the replica circuit 65 and the VCO. More generally,when a voltage level is switched between the replica circuit 65 and theVCO 12, the magnitude of the voltage spikes can be limited to a rangebetween zero and two percent of the operating voltage applied to theVCO.

The replica circuit 65 is a static DC circuit which tracks the impedancecharacteristics of a dynamic circuit such as the three stage inverterring oscillator of FIG. 1C. For a given bias and consequent frequencyresponse, f, the group delay, i.e., the collective phase shift, of thethree stages is 360°. Hence, this means each inverter stage has a delayof 120° and each of the nodes I1, I2, and I3, due to the phase shift,will be at a different potential (gate to ground) at any given time.With regard to the circuit 65 of FIG. 10, the device 102 is a PMOS FEThaving the gate connected to ground. This corresponds to one or more ofthe PMOS transistors in FIG. 1C when the potential of one of the gatenodes N₁, N₂, N₃ is close to zero. The device 108 is an NMOS FET havingthe gate connected to V_(REP). This represents one or more of the NMOStransistors in FIG. 1C when the potential of one of the nodes I₁/I₂/I₃is close to V_(OSC). The gate of the PMOS FET 104 is connected to a nodebetween the FETs 106 and 108, resulting in a gate potential which isneither low nor high such that the NMOS FET 104, connected to a nodebetween 102 and 104 that is neither at V_(REP) or ground. Thiscorresponds to the voltage level at one of the nodes N1, N2, N3 when thenode is in between a high state (V_(REP)) and a low state (ground). Tosummarize, the devices 102 and 108 operate in triode mode, whichcorresponds to the impedances of some of the transistors in the VCO ringthat are fully switched due to their gate biasing voltage being atV_(REP) or ground; while the devices 104 and 106 operate in thesaturation mode which corresponds to the impedances of some of thetransistors in the VCO ring oscillator that are partially switched ondue to their gates being biased at a voltage between V_(REP) and ground.

To summarize, the proportional path circuitry 60, 60′ receives inputsfrom the PFD 20 which generates pulses where the pulse width differenceis proportional to the phase difference of the feedback clock and thereference clock. The proportional path has three modes of operationbased on the signals UP, UN, DP, and DN. In FIG. 9A, the phase of thereference clock is shown to lead the feedback clock and there is apositive current injection into the VCO. In FIG. 9B, the phase of thereference clock is shown to lag the feedback clock. Hence there isnegative current injection into the VCO. FIG. 9C illustrates thecondition where there is no phase mismatch between the reference clockand the feedback clock. Hence, there is no change in the total netcurrent injected into the VCO.

In the past the charge pump, loop filter, and trans-conductanceamplifier have been necessary to translate the full swing up and downsignals from the PFD into the VCO as described with respect to FIG. 1B.Use of a loop filter having a large passive device is a growingimpediment in monolithic manufacturing processes that move past the 45nanometer node. Further, the loop filter provides a path for noisetransfer to the VCO. By eliminating the charge pump, loop filter, andthe small signal G_(m) amplifier, associated with the analog PLL ofFIGS. 1, the proportional path is simplified in accord with FIGS. 3, 6and 8, resulting in several advantages. The small signal G_(m) amplifieris a wide band device because it typically needs to process signals witha bandwidth up to several hundred megahertz. Hence, in prior PLLcircuits most of the noise from the current sources in the charge pump,as well as thermal noise from the resistor R of FIG. 1C have passeddirectly through the G_(m) amplifier into the VCO. Also, noise inherentin the G_(m) amplifier has been injected into the VCO. Consequently, thehigh gain path has led to greater noise amplification. Elimination ofthis block removes unnecessary noise which would otherwise be generatedfrom the charge pump, the loop filter, and the G_(m) amplifier. Instead,in the PLL circuit 10 the signals are sent directly from the PFD 20 tothe control and interface circuitry 22. The output of the PFD 20 simplycontrols switches, as described in FIGS. 6-9. This design eliminates apath for noise to travel to the VCO 12. Another feature of the PLLcircuit 10 is that the current source is heavily filtered, resulting ina reduced noise level. However, lack of gain using this method limitsthe system bandwidth. To introduce more gain, the DC current of theproportional path is made relatively high. In prior PLL circuits thiswould normally create a complication in device matching and repeatedgeneration of current spikes. Thus this architecture has not been usedpreviously due to these performance limitations. Now, by providing theseparate VCO replica circuit 65 as shown in FIG. 6, these issues areresolved. V_(REP) closely follows V_(OSC), allowing large currents to beswitched between the replica circuit 65 and the VCO 12 without creatinglarge current spikes.

With reference again to FIG. 6A, the circuitry 46 which forms the fastintegral path comprises a FET 122 connected between V_(DD) and the inputnode 45. The source terminal of the FET 122 is connected to V_(DD) andthe drain terminal of the FET 122 is connected to the node 45. The fastintegral path circuitry 46 is programmable and in the present embodimentconducts up to twenty percent of the total input current to the VCO 12,e.g., ten percent. The cut-off frequency of signals propagating throughthe fast integral path circuitry 46 is limited by the transconductanceof the FET 50 and the capacitor 58. The capacitance is selectable tolimit the amount of noise passing into the FET 122. The bandwidth of theDevice 122 is about ten MHz.

Still referring to FIG. 6A, the circuitry 48 which forms the slowintegral path comprises a FET 126 connected between V_(DD) and the inputnode 45. The source terminal of the FET 126 is connected to V_(DD) andthe drain terminal of the FET 126 is connected to the node 45. Thecircuitry 48 includes the low pass filter 49 through which the signalisum passes to remove high frequency noise before input to the gate 130of the FET 126. The bandwidth of the low pass filter 49 is programmable,i.e., adjustable, and may, for example, range from five KHz to one MHz,thus limiting the bandwidth of devices 126 and 68 to the same range.However, signal propagation through the proportional path circuitry iscontrolled by switches 64, 66 (see FIG. 6A), where device 68 providesthe bias current. The proportional path bandwidth is limited by theimpedance at the input node 45, which varies between 100 MHz and 400MHz.

The fast integral path (circuitry 46) and the slow integral circuit path(circuitry 48) control the frequency of the VCO 12. Changes inapplication environmental parameters, e.g., temperature and supplyvoltage, can affect the frequency of the VCO output signal 14. A changein the absolute temperature changes transistor switching speeds whichresult in a change in VCO frequency. The integral path compensates forthe VCO frequency variations due to these parameters and stabilizes theVCO frequency at the desired value. Parameters such as off chip supplyvoltage and temperature change at a very slow rate, typically atkilo-hertz rate or slower. Hence, the slow integral path is designed tooperate at a bandwidth as low as five KHz. However, other parameterssuch as reference clock frequency modulation can be set as high as 133KHz. The fast integral loop is designed to operate at a ten MHzbandwidth to ensure it can remove frequency errors due to referenceclock frequency modulation. In both cases, the bandwidth should behigher than the possible change rate in order to quickly correct fordeviations resulting from these and other environmental parameters.

To minimize jitter in the VCO 12 it is important to reduce or eliminatenoise from the power supply rails. In the past, linear regulators havebeen used to provide power supply rejection. However, this type ofimplementation requires feedback circuits and a reference voltage. Afeature of embodiments of the invention is that neither a feedbackcircuit nor a reference voltage is required to provide power supplyrejection. As shown generally in FIG. 6B, the phase locked loop circuit10 comprises Power Supply Rejection (PSR) subcircuitry 132 connected toprovide the supply voltage V_(DD) to other subcircuitry 133 of the phaselocked loop circuit 10, including the control and interface circuitry 22(e.g., fast integral path circuitry 46, control circuitry 47, slowintegral circuit path circuitry 48 and proportional path circuitry 60,60′). Example designs of the PSR subcircuitry 132 showman FIGS. 12-14,are referenced as subcircuitry 132 a, 132 b and 132 c.

With respect to transistor devices as now described in the context ofthe PSR subcircuitry, and other devices illustrated with reference tothe phase locked loop circuit 10, the disclosed embodiments incorporatefield effect transistors (FETs) but the invention is not so limited. Inthe context of embodiments which utilize FETs, the term region refers toa distinct and identifiable portion of a transistor, such as a source, adrain or a gate, and the term region may be used interchangeably withany one of these and interchangeably with a terminal to which one ofthese is in electrical conduction. The term source/drain region as usedherein means a semiconductor region or a terminal leading to thesemiconductor region wherein the region operates as a source or as adrain of a transistor device. Filter or filter element means one or morecapacitor devices, which are illustrated herein as two terminal devices.

A filter element may include one or more resistors, and reference to afilter element includes reference to impedance networks generally. Theterm filter refers to a filter which may be an analog filter or adigital filter. Low pass filter means a filter which is has a frequencycharacteristic wherein above a given frequency there is notable signalattenuation. Terminal and connection may refer to a point of contactwhich effects connection although in highly integrated circuitry aphysical connection may not be a characterized by a distinct connectionpoint which can be isolated from other conductive material. Also,reference to a point of connection or a terminal which receives a sourceof external power or voltage is to be understood as a point in circuitrywhich may receive such power or voltage during circuit operation butwhich may not be present when the circuitry is not operating.

The circuitry of FIG. 12 is exemplary of filter circuitry providingpower supply rejection for the exemplary phase locked loop subcircuitry133. PSR subcircuitry 132 a is connected to an exemplary terminal orconnection point 139 to receive a first supply voltage P_(v) from asource external to the phase locked loop circuit 10 and provide a secondsupply voltage, V_(DD), to the phase locked loop subcircuitry 133through an exemplary connection 137 b. The PSR subcircuitry 132 a isconfigured with the drain 136 d of a NMOS transistor 136 connected to anexemplary terminal 137 a at which the supply voltage P_(v) may beprovided from the external source. The source 136 s of the NMOStransistor 136 is connected to provide current to the phase locked loopsubcircuitry 133 through an exemplary connection 137 b. A low passfilter 138 is connected between the gate 136 g of the transistor 136 andthe exemplary terminal or connection point 139 to drive the highimpedance gate 136 g of the transistor with a filtered version of thesignal derived from the first supply voltage P_(v) when received fromthe external supply voltage source. Application of the filtered versionof the signal to the gate 136 g provides a voltage V_(DD) at the sourceterminal 137 b which exhibits power supply rejection above a cut-offfrequency set by the filter, i.e., determined in part by the filtercharacteristics. Below the cut-off frequency V_(DD) tracks P_(v). Abovethe cut-off frequency V_(DD) does not track variations in the supplyvoltage P_(v). In other embodiments, additional NMOS transistors and orfilters may be stacked to increase the amount of power supply rejection.See, for example, the circuits of FIGS. 12 and 13.

The circuitry of FIG. 13 includes PSR subcircuitry 132 b connected toreceive a first supply voltage P_(v) and provide a second supply voltageV_(DD) to the phase locked loop subcircuitry 133 through the exemplaryconnection 137 b. The PSR subcircuitry 132 b comprises aresistor-capacitor network and multiple PMOS transistors. In thisexample, two transistors 150, 152, two low pass filters 154, 156 and tworesistors 158, 160 are illustrated, it being understood that otherembodiments may comprise additional transistors, low pass filters andresistors. The transistors 150, 152 are arranged in series with thesource 150 s of the transistor 150 connected to an exemplary powersupply terminal 137 a at which a supply voltage P_(v) may be providedfrom a source external to the PLL circuit 10. The drain 150 d of thePMOS transistor 150 is connected to the source 152 s of the PMOStransistor 152. The drain 152 d of the PMOS transistor 152 is connectedto provide current to the phase locked loop subcircuitry 133 through theexemplary connection 137 b.

The high impedance gates 150 g 152 g of the two PMOS transistors 150,152 are each connected to an exemplary terminal or connection point 139,at which the supply voltage P_(v) may be received from a source externalto the PLL circuit 10. Resistors 158 and 160 are positioned betweenP_(v) and a reference terminal V_(SS). Each of the two low pass filters154, 156 is connected on a different side of the resistor 158 while theresistor 160 further limits current flow to V_(SS). The gate 150 g ofthe transistor 150 is connected through the low pass filter 154 isdriven with a signal derived from the first supply voltage P_(v) whenreceived from the external supply voltage source.

The gate 152 g of the transistor 152 is connected in series through thelow pass filter 154 and the resistor 158 to also receive and be drivenwith a signal derived from the first supply voltage P_(v), when receivedfrom the external supply voltage source. The filter 156 is connectedbetween the gate 152 g and a connection point 161 between the resistor158 and the resistor 160.

With this arrangement the drain 150 d of the first transistor 150provides a first modified limits current flow to V_(SS).

The filters 154, 156 provide the supply voltage V_(DD) at the terminal137 b which exhibits power supply rejection above a cut-off frequencyset by the filters 154, 156, i.e., determined in part by the filtercharacteristics. Below the cut-off frequency V_(DD) tracks P. Above thecut-off frequency V_(DD) does not track variations in the supply voltageP. In other embodiments, additional transistors and filters may beincorporated in the subcircuitry 132 b to increase the amount of powersupply rejection.

The circuitry of FIG. 14 includes PSR subcircuitry 132 c connected toreceive a first supply voltage P_(v) and provide a second supply voltageV_(DD) to the phase locked loop subcircuitry 133 through the exemplaryconnection 137 b. The PSR subcircuitry 132 c comprises aresistor-capacitor network and multiple PMOS transistors. In thisexample, two PMOS transistors 150, 152, two low pass filters 154, 156and two resistors 158, 160 are illustrated, it being understood thatother embodiments may comprise additional transistors, low pass filtersand resistors. The transistors 150, 152 are arranged in series with thesource 150 s of the transistor 150 connected to an exemplary powersupply terminal 137 a at which a supply voltage P_(v) may be providedfrom a source external to the PLL circuit 10. The drain 150 d of thePMOS transistor 150 is connected to the source 152 s of the PMOStransistor 152. The drain 152 d of the PMOS transistor 152 is connectedto provide current to the phase locked loop subcircuitry 133 through theexemplary connection 137 b. The high impedance gates 150 g 152 g of thetwo PMOS transistors 150, 152 are each connected to one or moreexemplary terminal or connection points 139, at which the supply voltageP_(v) may be received from a source external to the PLL circuit 10.Resistors 158 and 160 are positioned between P_(v) and a referenceterminal V_(SS).

The low pass filter 154 is connected at a node 161 between the resistors158, 160 so that the gate 150 g of the transistor 150 is connected inseries, through the filter 154 and the resistor 158, to be driven with asignal derived from the first supply voltage P_(v), when received fromthe external supply voltage source at a terminal or connection point139. The resistor 160 further limits current flow to V_(SS). The lowpass filter 156 is connected between the gate 152 g of the transistor152 and a terminal or connection point 139 to receive a signal derivedfrom the first supply voltage P_(v) when received from the externalsupply voltage source, and drive the gate 152 g of the transistor 152with the derived signal.

The filters 154, 156 provide the supply voltage V_(DD) at the terminal137 b which exhibits power supply rejection above a cut-off frequencyset by the filters 154, 156, i.e., determined in part by the filtercharacteristics. Below the cut-off frequency V_(DD) tracks P. Above thecut-off frequency V_(DD) does not track variations in the supply voltageP. In other embodiments additional PMOS transistors and filters may befurther incorporated to increase the amount of power supply rejection.

In the designs of FIGS. 12, 13 and 14, the transistors are kept insaturation to provide maximum power supply rejection, but they canoperate in other regions. With the filter techniques of FIGS. 12-14, theregulated supply voltage V_(DD) moves with respect to P_(v) up to thebandwidth of the low pass filter whereas, in previous designs, thesupply going to the VCO has been regulated via feedback to ensure thesupply stays constant.

In summary, the phase locked loop circuit 10 runs with lower jitter thana typical analog PLL, but contains a reduced number of analog blockscompared to an analog PLL. In an advantageous embodiment, the VCO 12 isan analog component while all other components are digital, thisrendering the design more suitable for low voltage operation and moreportable to current and future small geometry manufacturingtechnologies. PLL circuits according to the invention are also much lessprocess and environmentally sensitive than PLL analog designs. Theloop-dynamics of the PLL circuit 10 can be described by the cut-offfrequency:

F _(cut-off) =K _(VCO)/(2πN)  (1)

where K_(VCO) is the analog tuning gain of the VCO and N is the PLLfeedback divider value (DIV/N). As can be seen from Equation (1),K_(VCO) is the only process sensitive parameter in the loop dynamics ofthe PLL circuit 10. This results in relatively stable operation, makingthe PLL circuit 10 very robust and suitable for large volume production.

Numerous inventive features have been described. These include (1) phaselocked loop circuitry having a triple path for phase and frequencycontrol, there being a proportional path, a fast integral path and aslow integral path; (2) programmable control of the slow integral pathby adjusting its bandwidth for use in loop dynamics; and (3) an openregulation technique which does not use feedback to achieve power supplyrejection. The proportional path eliminates the need for a charge pump,loop filter, or equivalent digital PLL techniques by its direct usage inswitching current up to 40% of the total VCO bias current. An embodimentuses a filtered clean bias voltage from the slow integral path to biasthe proportional path current. Also, a new static no feedback replicacircuit is disclosed which tracks a dynamic circuit's average switchingcurrent. These and other concepts are disclosed herein for numerousapplications and the embodiments and specific applications shown hereinare not to be construed as limiting.

The inventive concepts may be advantageously applied in a variety ofelectronic systems. As one example, there is a need for increased ratesof data transfers between devices, e.g., integrated circuits and, for avariety of reasons, these transfers are performed with high speedserializer/deserializer devices commonly referred to as Serves. In lieuof having n-bit wide parallel data transfers between devices, data isserialized to reduce the level of parallelism. This reduces the pincount of each device, but the data transfer frequency increases ininverse proportion to the ratio by which the pin count is reduced.Typically, to control the movement of data transfers, each deviceestablishes a data transfer rate by multiplying a reference clock speedwith intermediate frequency PLL circuitry. The increased clock speed maybe further stepped up with one or more other PLL circuits in each of oneor more Serves cores formed on the device to control specific dataclocking functions. FIG. 15 is a simplified schematic illustrating aserializer 162 from which data is transferred in a first device 164(e.g., a first integrated circuit) to a deserializer 166 in a seconddevice 168 (e.g., a second integrated circuit). The first device 164 isa component of a first system 170 and may be mounted on a first pcboard. The second device 168 is a component of a second system 171 andmay be mounted on a second pc board. The serializer 162 receives m-bitwide parallel data from circuitry 167 within the first device 164 andconverts the data to a lower bit width, k, of parallel data for transferalong the serialized data line 172 to the deserializer 166 which thenperforms a post transfer restoration of the data to the original m-bitwidth for operation thereon by other circuitry in the second device 168.The serializer 162 and the deserializer 166 each include one or morephase locked loop circuits 10 which receive a reference clock signal tofacilitate operation of first control circuitry 173 of the serializer162 or operation of second control circuitry 175 of the deserializer166. The m-bit wide data is initially received by first input circuitry174 of the serializer, then undergoes parallel-to-serial conversion inserialization block 176, from which the data passes through first outputcircuitry 178 and are then transmitted off chip via the data line 172.The serialized data stream is received by the second input circuitry 184of the deserializer 166 and then undergoes serial-to-parallel conversionin deserialization block 186 to recreate m-bit wide parallel data whichpass through second output circuitry 188 to other circuitry 190 in thesecond device 168 for processing. The phase locked loop circuits 10illustrated in FIG. 15 are coupled to control circuitry 173 or 175 toprovide timing and control which assures stability of the timing signalsaccording to which data is serialized, transferred and desterilized.Thus the inventive concepts may be implemented on two separate systems170 and 171, each having a device comprising circuitry (I) forperforming serialization of data from a m-bit wide parallel arrangementto a k-bit wide parallel arrangement for transfer of the data to anotherdevice, or (ii) for performing deserialization of data from a k-bit wideparallel arrangement to a m-bit wide parallel arrangement for transferof the data to another device, where k<m.

While the invention has been described with reference to particularembodiments, it will be understood by those skilled in the art thatnumerous inventive concepts disclosed herein can be implemented in avariety of circuit applications and systems. Many of the aforedescribedimprovements can, for example, be implemented in delay locked loop (DLL)circuitry to adjust the phase of a signal or for clock recovery.Although not illustrated, with reference generally to the foregoingfigures and, specifically, the phase locked loop circuitry 10, it willbe understood by those skilled in the art that such DLL circuitryaccording to the invention will comprise phase detector circuitry inlieu of the illustrated phase and frequency detector 20, and a chain ofdelay gates in lieu of the VCO 12. DLL circuitry incorporating featuresof the invention may be integrated into, for example, memory devicessuch as Dynamic Random Access Memory (DRAM) devices.

Further, various modification to the described embodiments arecontemplated and equivalents may be substituted for elements thereofwithout departing from the spirit of the invention. Accordingly, thescope of the invention is only limited by the claims which follow.

1. A method of changing the phase or frequency of a first periodicsignal output from a first nonlinear device having a nonlinear impedancecharacteristic with respect to a second periodic signal, the devicehaving a voltage operating range, the method comprising: providingdiscrete control signal types of variable time width in proportion to aphase difference between the first and second signals, a first of thesignal types indicative of a negative difference in phase or frequencyof the first periodic signal relative to the phase or frequency of thesecond periodic signal, and a second of the signal types indicative of apositive difference in phase or frequency of the first periodic signalrelative to the phase or frequency of the second periodic signal;periodically applying control signals of the first type to a firstswitching device to control current flow along a first circuit path froma voltage source through the first switching device, through the firstnonlinear device and to a reference voltage terminal to adjust a phaseor frequency characteristic of the first signal output from the firstnonlinear device; and periodically applying signals of the second typeto second switching circuitry to control current flow along a secondcircuit path in parallel with the first circuit path from the voltagesource through the second switching device, through a second impedancedevice having a nonlinear impedance characteristic and to the referencevoltage terminal, wherein the first and second switching devices areoperated in to response to variations in the signals of the first andsecond types to switch current flow between the first and second circuitpaths so that at times current flows only through the first circuit pathand then only through the second circuit path, wherein: the system doesnot incorporate charge pump circuitry to store charge in a capacitor forinput to the voltage controlled oscillator; and impedancecharacteristics of the first nonlinear device and the second nonlinearimpedance device are so matched that current-voltage characteristics ofthe first and second impedance devices are within five percent of oneanother throughout the voltage operating range of the VCO.
 2. The methodof claim 1 wherein, to change the phase or frequency of the firstperiodic signal, with application of the signal types to the switchingdevices: (i) during a portion Δt of a period of the first or secondperiodic signal, one of the first and second switching devices is in aconduction mode while the other switching device is not in a conductionmode, thereby providing current flow only through the first impedancedevice in the first branch or providing current flow only through thesecond impedance device in the second branch or providing no currentflow through the first impedance device and no current flow through thesecond impedance device.
 3. The method of claim 1 wherein the firstimpedance device is a voltage controlled oscillator.
 4. A systemcomprising proportional path circuitry for use in changing the phase orfrequency of a first periodic signal with respect to a second periodicsignal based on an output from a comparator, comprising: a detector fordetermining a phase or frequency difference between the first and secondsignals, which provides discrete periodic signal types of variable timewidth in proportion to a phase difference between the first and secondsignals, a first of the signal types indicative of a negative differencein phase or frequency of the first periodic signal relative to the phaseor frequency of the second periodic signal, and a second of the signaltypes indicative of a positive difference in phase or frequency of thefirst periodic signal relative to the phase or frequency of the secondperiodic signal; switching circuitry including first and second parallelbranches, each branch connected between a supply voltage connection,VDD, and a reference voltage connection, VSS, each branch including afirst switching device and a load device, the switching device of thefirst branch and the switching device of the second branch each coupledto receive a different one of the signal types as an input signal toplace the switching device in a mode to conduct current between thesupply voltage connection and the reference voltage connection or in anon-conducting mode wherein, during circuit operation, to change thephase or frequency of the first periodic signal, with application of thesignal types to the switching devices: during a portion Δt of a periodof the first or second periodic signal, one of the first and secondswitching devices is in a conduction mode while the other switchingdevice is not in a conduction mode, thereby providing current flow onlythrough the load device of the first branch or providing current flowonly through the load device of the second branch or providing nocurrent flow through either load device.
 5. The proportional pathcircuitry of claim 4 wherein said portion Δt of the period is based on adifference in time between when the first periodic signal transitionsfrom a first voltage level to a second voltage level and when the secondperiodic signal transitions from the first voltage level to the secondvoltage level.
 6. The proportional path circuitry of claim 5 wherein thefirst voltage level is a low voltage level and the second voltage levelis a high voltage level.
 7. The system of claim 4 wherein, duringcircuit operation, provision of the first of the signal types isindicative of increasing the phase or frequency of the first periodicsignal relative to the phase or frequency of the second periodic signal,and provision of the second of the signal types is indicative ofdecreasing the phase or frequency of the first periodic signal relativeto the phase or frequency of the second periodic signal.
 8. The systemof claim 4 wherein, during operation of the circuitry during a portionof said period of the first or second periodic signal other than duringsaid portion Δt of said period, the switching devices provide a steadystate current flow through one of the load devices.
 9. The system ofclaim 4 wherein, during operation of the circuitry during a portion ofsaid period of the first or second periodic signal other than duringsaid portion Δt of said period, the switching devices provide a steadystate current flow through both load devices.
 10. The system of claim 4wherein: each of the first and second branches includes a secondswitching device in parallel with the first switching device positionedin the same branch; and each of the second switching devices is underthe control of a signal type which is the inverse of the signal typeapplied to the first switching device of the same branch.
 11. The systemof claim 10 wherein: during the portion of the period during whichsteady state current flows one switch in each branch provides conductionto one of the load devices and the other switch is in a non-conductingmode.
 12. The system of claim 4 wherein the load device of the firstbranch is a voltage controlled oscillator and the load device of thesecond branch is a subcircuit having impedance characteristics whichsubstantially match impedance characteristics of the voltage controlledoscillator and the system does not incorporate charge pump circuitry tostore charge in a capacitor for input to the voltage controlledoscillator.
 13. The system of claim 4 wherein: the load device of thefirst branch is a voltage controlled oscillator and the load device ofthe second branch is a subcircuit having impedance characteristics whichsubstantially match impedance characteristics of the voltage controlledoscillator; and the impedance characteristics of the subcircuit soclosely follow the impedance characteristics of the voltage controlledoscillator as a function of voltage level as to allow a voltage level tobe switched between the subcircuit and the voltage controlled oscillatorwithout creating voltage spikes larger than five percent of the voltagelevel input to the voltage controlled oscillator.
 14. The system ofclaim 13 wherein the impedance characteristics of the subcircuit soclosely follow the impedance characteristics of the voltage controlledoscillator as a function of voltage level as to allow a voltage level tobe switched between the subcircuit and the voltage controlled oscillatorwithout creating voltage spikes larger than two percent of the voltagelevel input to the voltage controlled oscillator.
 15. The system ofclaim 13 wherein the impedance characteristics of the subcircuit soclosely follow the impedance characteristics of the voltage controlledoscillator as a function of voltage level as to allow a voltage level tobe switched between the subcircuit and the voltage controlled oscillatorwithout creating voltage spikes larger than one percent of the voltagelevel input to the voltage controlled oscillator.
 16. The system ofclaim 15 wherein the impedance characteristics of the subcircuit soclosely follow the impedance characteristics of the voltage controlledoscillator as a function of voltage level as to prevent creation of anyvoltage spikes when a voltage level is switched between the subcircuitand the voltage controlled oscillator.
 17. The system of claim 1 whereinthe first load device is a three stage inverter ring oscillator and thesecond load device is a subcircuit having characteristics which trackthe impedance characteristics of the oscillator.
 18. A circuit suitablefor tracking impedance characteristics of an oscillator comprising afirst PMOS FET, a second PMOS FET, a first NMOS FET and a second NMOSFET wherein: the first PMOS FET has a gate region connected to receive areference voltage; the second NMOS FET has a gate region connected toreceive a supply voltage level; the second PMOS FET has a gate regionconnected to a node between the first and second NMOS FETs; and thefirst NMOS FET has a gate region connected to a node between the firstand second PMOS FETs.
 19. The circuit of claim 18 wherein the oscillatoris a three stage inverter ring oscillator and the first PMOS FET and thesecond NMOS FET operate in a triode mode, which corresponds to theimpedances of FET transistors in the oscillator that are fully switchedwhen associated gate bias voltage is at the supply voltage level or thereference level (ground) while the second PMOS FET and the first NMOSFET operate in the saturation mode which corresponds to the impedancesof FET transistors in the oscillator that are partially switched on whenassociated gates are biased at a voltage between the supply voltagelevel and the reference level.